Power factor correction circuit

ABSTRACT

A power factor correction circuit structure is described. The circuit connects a bridge rectifier and a first capacitor in series. The first capacitor, a winding and a first switching device are also connected in series. The first switching device is the low-side switching device in a bridge converter. The first switching device, a second switching device and a second capacitor are also connected in series. The second switching device is the high-side switching device in the bridge converter and the second capacitor is the boost capacitor in the PFC circuit. The winding can be one additional winding of the main transformer in the bridge converter or an independent inductor.

FIELD OF THE INVENTION

[0001] The present invention relates to a power factor correction circuit and more particularly, to a power factor correction circuit for improving a power factor of a switching power supply designed in bridge topologies in order to comply with the requirements of Class A or Class D as stipulated in harmonic current rules IEC-1000-3-2.

BACKGROUND OF THE INVENTION

[0002] A typical switching power supply is shown in FIG. 1. The supply comprises an AC/DC rectifier 1, and a DC/DC converter 2 in which an electrolytic capacitor C₃ is connected as a filter for the bridge rectifier BD₁.

[0003]FIG. 5 discloses a circuit structure in which the DC\DC converter 2 shown in FIG. 1 is a half-bridge converter. In accordance with the design structure shown in FIG. 5, a bridge rectifier BD₁ is used to rectify the AC power V_(S1). A capacitor C₃ is then used to filter the rectified power and generates a DC voltage V_(C3). The capacitor C₁ and the capacitor C₂ are connected in a common node to form a voltage divider. Therefore, the voltage in the common node between the two capacitors is V_(C3)/2.

[0004]FIG. 7 is a time chart for the PWM signals, V_(HG) and V_(LG), which are driving signals for the switch Q₁ and the switch Q₂ shown in FIG. 5, respectively. Both PWM signals V_(HG) and V_(LG) are low (low voltage) for 0≦t≦t1. At this time, the switch Q₁ and the switch Q₂ are both turned off. Therefore, the output voltage V_(O) is supplied by the capacitor C₄.

[0005] When t1≦t≦t2, the PWM signal V_(LG) is high (high voltage) and the PWM signal V_(HG) is low (low voltage). At this time, the switch Q₁ is turned off and the switch Q₂ is turned on. The current flows through the capacitor C₁, the primary winding P₁ and the switch Q₂ to the ground. Under this situation, the transformer transfers the power from the primary winding P₁ to the secondary winding S₁ to supply power to the capacitor C₄ and output a voltage V_(O).

[0006] When t2≦t≦t3, the switch Q₁ and the switch Q₂ are both turned off. At this time, the operation state of the circuit is same as the operation state of the circuit at 0≦t≦t1.

[0007] When t3≦t≦t4, the PWM signal V_(LG) is low (low voltage) and the PWM signal V_(HG) is high (high voltage). At this time, the switch Q₁ is turned on and the switch Q₂ is turned off. Therefore, the current flows through the switch Q₁, the primary winding P₁ and the capacitor C₂ to the ground. The transformer transfers the power from the primary winding P₁ to the secondary winding S₂ to supply power to the capacitor C₄ and output a voltage V_(O).

[0008] The power switching cycle described above is then performed repeatedly to supply power to a loading. On the other hand, the output voltage V_(O) is transferred to a feedback system 12. The feedback system 12 feeds a signal back to the high-frequency pulse signal control circuit 50 to modify the duty cycle of the PWM signals V_(HG) and V_(LG). For example, if the power supplied to the load is insufficient when the output voltage is lower than a required value, the feedback signal enlarges the duty cycle of the PWM signals V_(HG) and V_(LG) to increase the conduction time of the switch Q₁ and the switch. Q₂. Therefore, the time for transferring power from the primary winding to the secondary winding of the transformer T₁ is increased. In other words, the power supplied to the secondary winding is increased. The output voltage V_(O) is therefore also increased. Finally, the output voltage V_(O) again attains the required voltage. This means, however, that the power supplied to the load is overdriven when the output voltage is higher than the required value. In this situation, the duty cycle of the PWM signals V_(HG) and V_(LG) should be reduced.

[0009] Note that the input current I_(pc) in FIG. 5 is a pulse current as shown in the graph of FIG. 2. The power factor of the conventional switching power supply is significantly decreased (e.g., approximately 50%) due to the distorted input current, and the total harmonics distortion (hereinafter referred as THD) is even higher than 100% after the rectification performed by the AC/DC rectifier 1 shown in FIG. 1. As a result, the total harmonics is seriously distorted, the quality is poor, and, even worse, precious energy is wasted.

[0010] Thus, many countries have promulgated a number of harmonic current rules (e.g., IEC-1000-3-2) which specify the current wave shape of the power supply for manufacturers to obey in order to improve the efficiency and quality of the power source being supplied.

[0011] As such, various designs of power factor correction circuits have been proposed by researchers in order to improve power factor of the conventional switching power supply. Two examples of typical prior art are described in the following:

1. Inductor Type Power Factor Correction Circuit

[0012] As shown in FIG. 3, the prior art discloses a design in which a low frequency large winding L₁ is in series between a bridge rectifier BD₁ and an electrolytic capacitor C₁. The winding L₁ and the capacitor C₁ form a low pass filter to rectify the input current of a DC/DC converter 2. Such design is similar in function to the ballast for correcting the power factor of a fluorescent lamp. However, winding L₁ relatively large, has only a limited power factor improvement, and creates an abnormally high temperature during operation.

2. Active Type Power Factor Correction Circuit

[0013] As shown in FIG. 4, the prior art discloses a design in which the AC/DC rectifier is redesigned to form a two-stage circuit with the DC/DC converter 2. Further, a complex control circuit 11 and a large switch element Q₁ are added therein to improve the power factor, However, it is relatively complex in circuit design and is expensive to manufacture,

[0014] Many power factor correction circuits have been developed based on the basic concepts involved in the two examples of prior art mentioned above, and with similar drawbacks.

SUMMARY OF THE INVENTION

[0015] In accordance with the foregoing description, there are many drawbacks in the conventional power factor correction circuit. For example, the circuit structure depicted in FIG. 3 is relatively large, while the circuit structure depicted in FIG. 4 is relatively complex in circuit design and is expensive to manufacture.

[0016] Therefore, the main purpose of the present invention is to provide a power factor correction circuit with a high power factor.

[0017] Another purpose of the present invention is to provide a power factor correction circuit to solve the problems existing in the prior art.

[0018] A further purpose of the present invention is to provide a switching power supply structure that is small and economical to manufacture. It is an object of the present invention to provide a power factor correction circuit comprising a series connection of a bridge rectifier and a first capacitor. The first capacitor, a winding and a first switching device are connected in series. The first switching device is the low-side switching device in a bridge converter and is connected with a first anti-parallel diode. The first switching device, a second switching device and a second capacitor are also connected in series. The second switching device is the high-side switching device in the bridge converter and is connected with a second anti-parallel diode. The second capacitor acts as a boost capacitor in the PFC circuit and provides the DC operating voltage for the bridge converter. The winding can be one additional winding of the main transformer in the bridge converter or an independent inductor. Further, the power factor of the off-line switching power supply is increased to above 0.9 by appropriately selecting the value of the first capacitor and the winding in order to comply with the requirements of Class A or Class D as stipulated in harmonic current rules IEC-1000-3-2. Furthermore, the inserted PFC circuit does not affect the normal operation of the bridge converter.

BRIEF DESCRIPTION OF THE DRAWINGS

[0019] The foregoing aspects and many of the attendant advantages of this invention will become more readily appreciated and better understood by referencing the following detailed description, when taken in conjunction with the accompanying drawings, wherein:

[0020]FIG. 1 is a circuit diagram of a prior art off-line switching power supply;

[0021]FIG. 2 is a graph showing the wave shapes of the input voltage versus the input current of FIG. 1;

[0022]FIG. 3 is a circuit diagram of an inductor type power factor correction circuit;

[0023]FIG. 4 is a circuit diagram of an active type power factor correction circuit;

[0024]FIG. 5 is a circuit diagram of a switching power supply designed in half-bridge topology;

[0025]FIG. 6a is a circuit diagram of a preferred embodiment of the present invention;

[0026]FIG. 6b is a effective circuit diagram of a preferred embodiment of the present invention, wherein the transformer T1 shown in FIG. 6a is replaced by an ideal effective transformer model;

[0027]FIG. 7 is a timing diagram of the PWM signals, V_(HG) and V_(LG), which are usually used to drive the switching devices in conventional bridge converters;

[0028]FIG. 8 is a simplified circuit diagram of the primary-side circuit in FIG. 6b operating in the durations 0<t<t1 and t2<t<t3 according to FIG. 7;

[0029]FIG. 9 is a simplified circuit diagram of the primary-side circuit in FIG. 6b operating in the duration t1<t<t2 according to FIG. 7;

[0030]FIG. 10 is a simplified circuit diagram of the primary-side circuit in FIG. 6b operating in the duration t3<t<t4 according to FIG. 7;

[0031]FIG. 11a is a graph showing the waveforms of the critical voltages in FIG. 6b in order to describe the power factor correction ability of the present invention;

[0032]FIG. 11b is a graph showing the waveforms of the input voltage and current in the embodiment shown in FIG. 6b; and

[0033]FIG. 12 is a circuit diagram of a second embodiment of the present invention;

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

[0034] Without limiting the spirit and scope of the present invention, the circuit structure proposed in the present invention is illustrated with one preferred embodiment. One with ordinally skill in the art, upon acknowledging the embodiment, can apply the circuit structure of the present invention to various switching power supply. The circuit structure of the present invention allows high power factor and relative small volume. Additionally, the present invention does not require an additional inductor. Therefore, the size of the circuit structure is reduced and the manufacturing cost is also reduced. The application of the present invention is not limited by the preferred embodiments described in the following.

[0035] The present invention provides a circuit structure including a power factor correction circuit and a switching power supply.

[0036]FIG. 6a depicts the first preferred embodiment of the present invention. Compared to FIG. 5, the transformer T₁ has additional primary windings P₂ and P₃. In FIG. 6a, the voltage V_(S1) is the AC input power. The current I_(S1) is the current generated by the AC input power. The bridge rectifier BD₁ and the capacitor C₃ rectify the AC power to DC power. The high-frequency pulse signal control IC outputs the PWM signals V_(HG) and V_(LG) for controlling the state of the switch Q₁ and the switch Q₂. Two diodes D₁ and D₂ respectively connected in anti-parallel with the switch Q₁ and the switch Q₂. The feedback system receives the output signal, V_(O), transferred from the output end and sends out a feedback signal to the high-frequency pulse signal control IC. Then, the high-frequency pulse signal control IC modulates the duty cycle of the PWM signals V_(HG) and V_(LG) to steady the output signal V_(O).

[0037]FIG. 6b replaces the transformer T₁ depicted in FIG. 6a with an ideal transformer model. The windings P₁, P₂, P₃, S₁ and S₂ all are ideal coupling windings. L_(m) is the effective magnetizing inductance referred to the P₃ side.

[0038] Referring to FIG. 6b, the transformer T₁ has an additional winding pair P₂ and P₃. L_(m) is the effective magnetizing inductance referred to the P₃ side. On the other hand, a boost capacitor C_(boost) is used in the circuit structure. Therefore, the switch Q₂, the diode D₁, the boost capacitor C_(boost) and the inductor L_(m) compose a boost circuit to perform a power factor correction function. It is noted that the waveform of the voltage V_(C3) has to be a full wave to increase the efficiency of the power factor correction function and the lowest voltage value of the voltage V_(C3) has to be higher than zero. Therefore, the capacitance of the capacitor C₃ cannot be too large. The resistor R₁, the capacitor C₅, the diode D₃ and the winding P₂ compose an auxiliary power supply to provide power for the high-frequency pulse signal control IC. Although the auxiliary power supply is not shown in FIG. 5, it is a common design in a conventional converter and, of course, it is not a block of the PFC circuit according to the present invention. The stead-state operating condition of FIG. 6b is described in the following paragraphs.

[0039] Under steady-state operating conditions, the boost circuit composed of the switch Q₂, the diode D₁, the boost capacitor C_(boost) and the inductor L_(m) make a voltage drop V_(boost) across the capacitor C_(boost) higher than the V_(C3).

[0040]FIG. 7 shows a time chart of the PWM signals V_(HG) and V_(LG) for driving the switch Q₁ and the switch Q₂, respectively. At 0≦t≦t₁, the PWM signal V_(HG) and the PWM signal V_(LG) both are low (low voltage). Therefore, the switch Q₁ and the switch Q₂ both are turned off. The diode D₁ is forward biased. At this time, the voltage drop V_(P3) across the winding P₃ is shown as follows:

V _(P3) =V _(C3) −V _(boost)

[0041] The voltage value V_(P3) is less than zero because the voltage value V_(boost) is larger than the voltage value V_(C3). The current I_(L) flowing through the inductor L_(m) decreases gradually. And, the capacitor C₁, the capacitor C₂ and the capacitor C_(boost) are charged by the magnetizing inductor L_(m). At this time, no power is transferred to the load (not shown) by the transformer T₁. Therefore, the power required by the load is supplied by the capacitor C₄.

[0042] Referring to FIG. 8, a simplified circuit diagram of the primary-side circuit is shown without the part of the auxiliary power supply. The dotted line in FIG. 8 indicates the flowing direction for current I_(L) at 0≦t≦t₁. It is noted that the transformer T₁ is simplified to an ideal model. The symbol L_(m) means the magnetizing inductor and the windings P₁, P₂, P₃, S₁ and S₂ shown in the FIG. 6b only perform a coupling function,

[0043] At t₁≦t≦t₂, the PWM signal V_(LG) is high (high voltage) and the PWM signal V_(HG) is low (low voltage). At this time, the switch Q₁ is turned off and the switch Q₂ is turned on. The diode D₁ is reverse biased. At this time, the voltage drop V_(P3) across the winding P₃ is shown as follows:

V_(P3)=V_(C3)

[0044] The voltage V_(C3) is larger than zero. Therefore, the current I_(L) flowing through the inductor L_(m) increases gradually. Therefore, the voltage V_(C3) charges the inductor L_(m). On the other hand, the capacitor C_(boost) is discharged through the capacitor C₁, the winding P₁ and the switch Q₂. The capacitor C₂ is also discharged through the winding P₁ and the switch Q₂. At this time, the power is transferred from the primary winding P₁ to the secondary winding S₁. Then, the power is supplied to the load (not shown) and the capacitor C₄ from the secondary winding S₁.

[0045] Referring to FIG. 9, a simplified circuit diagram of the primary-side circuit is shown without the part of the auxiliary power supply. The dotted line depicted in FIG. 9 indicates the flowing direction for current I_(L) at t₁≦t≦t₂.

[0046] At t₂≦t≦t₃, the PWM signal V_(HG) and the PWM signal V_(LG) both are low (low voltage). Therefore, the switch Q₁ and the switch Q₂ are both turned off. The diode D₁ is turned on. At this time, the voltage drop V_(P3) across the winding P₃ is shown as follows:

V _(P3) =V _(C3) −V _(boost)

[0047] The voltage value V_(P3) is less than zero because the voltage value V_(boost) is larger than the voltage value V_(C3). The current I_(L) flowing through the inductor L_(m) decreases gradually. At this time, the operating condition of the circuit is the same as that of the circuit at 0≦t≦t₁. The simplified circuit diagram is also same as FIG. 8.

[0048] At t₃≦t≦t₄, the PWM signal V_(LG) is low (low voltage) and the PWM signal V_(HG) is high (high voltage). The switch Q₁ is turned on and the switch Q₂ is turned off. The diode D₁ is reverse biased. At this time, the voltage drop V_(P3) across the primary winding P₃ is shown as follows:

V _(P3) =V _(C3) −V _(boost)

[0049] The voltage value V_(P3) is less than zero because the voltage value V_(boost) is larger than the voltage value V_(C3). The current I_(L) flowing through the inductor L_(m) decreases gradually. Therefore, the magnetizing inductor L_(m) continues discharging through the primary winding P₁ and the capacitor C₂. On the other hand, the capacitor C_(boost) and the capacitor C₁ are both discharged through the switch Q₁ and the primary winding P₁. During this time, the capacitor C₂ is charged and power is transferred from the primary winding P₁ to the secondary winding S₂. Then, the power is supplied to the load (not shown) and the capacitor C₄ from the secondary winding S₂.

[0050] Referring to FIG. 10, a simplified circuit diagram of the primary-side circuit without the part of the auxiliary power supply is shown. The dotted line depicted in FIG. 10 indicates the flowing direction for current I_(L) at t₃≦t≦t₄.

[0051] Then, the power switching cycle (0≦t≦t₄) described in the above is repeated to supply power to a load. On the other hand, the output voltage V_(O) is transferred to a feedback system. The feedback system feeds a signal back to the high-frequency pulse signal control IC to modify the duty cycle of the PWM signals V_(HG) and V_(LG). For example, if the power supplied to the load is insufficient when the output voltage is lower than the specific value, the feedback signal enlarges the duty cycle of the PWM signals V_(HG) and V_(LG) to increase the conduction tire of the switch Q₁ and the switch Q₂. Therefore, the time for transferring power from the primary winding to the secondary winding of the transformer T₁ is increased. In other words, the power supplied to the secondary winding is increased. Therefore, the output voltage V_(O) is also increased. Finally, the output voltage V_(O) again attains the specific value. However, the power supplied to the load is overdriven when the output voltage is higher than the specific value. In this situation, the duty cycle of the PWM signals V_(HG) and V_(LG) should be reduced.

[0052]FIG. 11a is a graph showing the waveforms of the critical voltages, V_(C3), V_(boost), V_(S1) and V_(boost)−V_(C3), in FIG. 6b in order to describe the power factor correction ability of the present invention. In accordance with the above description and the time chart depicted in FIG. 7, the current I_(L) flowing through the inductor L_(m) increases gradually only when the switch Q₂ is turned on at t₁≦t≦t₂ because the voltage value V_(P3) is equal to V_(C3) and V_(C3) is larger than zero. At other times, 0≦t≦t₁, t₂≦t≦t₃ and t₃≦t≦t₄, the current I_(L) flowing through the inductor L_(m) decreases gradually because the voltage value V_(P3) is equal to V_(C3)−V_(boost) and V_(C3)−V_(boost) is less than zero.

[0053]FIG. 11b is a graph showing the waveforms of the input voltage and current in the embodiment shown in FIG. 6b. The waveform of the input current I_(S1) can be derived from the following description according to the timing diagram shown in FIG. 7. It is noted that the input current I_(S1) is approximately equal to the current I_(L), because the capacitance of C₃ is small enough. If Δt₁=t₂−t₁,

[0054] then, at Δt₁, switch Q₂ is turned on and the voltage value V_(P3) is eaqual to V_(C3).

[0055] And if Δt₂=t₄−Δt₁,

[0056] then, at Δt₂, the voltage value V_(P3) is eaqual to (V_(C3)−V_(boost)).

[0057] In accordance with the following equation for inductor: $\begin{matrix} {{V = {L\frac{i}{t}}}{{i} = \frac{V \cdot {t}}{L}}} & (1) \end{matrix}$

[0058] When the value of Δt and ΔIL is almost zero, equation (1) becomes the following equation: $\begin{matrix} {{\Delta \quad I_{L}} = \frac{{V \cdot \Delta}\quad t}{L_{m}}} & (2) \end{matrix}$

[0059] When the voltage value of V_(C3) is at the lowest level, the voltage value of (V_(boost)−V_(C3)) is larger than the V_(C3) in accordance with FIG. 11a. On the other hand, the value of Δt₂ is also larger than the value of Δt₁. Therefore, the following eaquation can be derived: $\begin{matrix} {\left( {V_{C3} \times \Delta \quad t_{1}} \right) < \left\lbrack {\left( {V_{boost} - V_{C3}} \right) \times \Delta \quad t_{2}} \right\rbrack} & (3) \\ {\frac{{V_{C3} \cdot \Delta}\quad t_{1}}{L_{m}} < \frac{{\left( {V_{boost} - V_{C3}} \right) \cdot \Delta}\quad t_{2}}{L_{m}}} & (4) \end{matrix}$

[0060] L_(m) is a constant. In accordance with equation (4), the current difference at Δt₁ is less than the current difference at Δt₂, but these two values are getting closer as VC₃ rising. Therefore, the current I_(S1) presents a vibration curve around zero ampere when the voltage V_(S1) is rising from zero voltage as shown in FIG. 11b.

[0061] Then, the current difference at Δt₁ is larger than the current difference at Δt₂ after the following equation (5) is derived:

(V _(C3) ×Δt ₁)=[(V _(boost) −V _(C3))×Δt ₂]  (5)

[0062] Therefore, the current I_(S1) begins to rise as shown in FIG. 11b.

[0063] After the peak value of V_(C3) is arrived, the value of V_(C3) starts to fall and the value of (_(V) _(boost)−V_(C3)) starts to rise. At this time, the value of the (V_(C3)×Δt₁) is still larger than the value of the (V_(boost)−V_(C3))×Δt₂. Therefore, the current I_(S1) still rises as shown in FIG. 11b.

[0064] In the situation where the value of (V_(C3)×Δt₁) is again equal to the value of the (V_(boost)−V_(C3))×Δt₂, the current difference at Δt₁ is less than the current difference at Δt₂ again. Therefore, the current I_(S1) begins to decrease as shown in FIG. 11b. Compared to FIG. 2, the waveform of the input current in FIG. 11b is an improvement.

[0065] In accordance with the above analysis, the switch Q₂, the diode D₁, the boost capacitor C_(boost) and the inductor L_(m) together compose a boost circuit to perform a power factor correction function. The operation method is similar to the operation method for the boost circuit shown in FIG. 4. Therefore, the present invention raises the power factor and the operation of the bridge converter is not disturbed. In accordance with the prior art depicted in FIG. 4, an additional control circuit is required to control the switch Q₁. However, in the present invention, the switch Q₂ is controlled by the high-frequency pulse signal control IC which belongs to the half-bridge converter. In other words, the present invention does not require an additional control circuit. On the other hand, the present invention also does not require the switch Q₁ shown in FIG. 4. Therefore, the present invention reduces the manufacturing costs.

[0066] The main difference between the half-bridge converter and the full-bridge converter is that the capacitor C₁ and the capacitor C₂ shown in FIG. 5 are replaced by the switch devices in the full-bridge converter. Two diodes respectively connected in anti-parallel with these two switches. Therefore, the circuit structure of the present invention may also be used in the full-bridge converter. The operation method is similar to that described above.

[0067] On the other hand, referring to FIG. 12, a circuit structure of another embodiment of the present invention is shown. Because the present invention only utilizes the magnetizing inductor of the winding P₃ shown in FIG. 6a, the winding P₃ can be replaced by an inductor as shown in FIG. 12. The operation method of the circuit structure shown in FIG. 12 is the same as the operation method of the circuit shown in FIG. 6a. Similarly, a full-bridge converter may also be used to replace the half-bridge converter shown in FIG. 12.

[0068] It is noted that some half-bridge converters (e.g., resonant half-bridge converters) do not include the upper capacitor C_(l). However, this kind of half-bridge topology may also be used in the present invention.

[0069] As is understood by a person skilled in the art, the foregoing descriptions of the preferred embodiment of the present invention are an illustration of the present invention rather than a limitation thereof. Various modifications and similar arrangements are included within the spirit and scope of the appended claims. The scope of the claims should be accorded to the broadest interpretation so as to encompass all such modifications and similar structures. While a preferred embodiment of the invention has been illustrated and described, it will be appreciated that various changes can be made therein without departing from the spirit and scope of the invention. 

What is claimed is:
 1. A power factor correction circuit for improving a power factor of a switching power supply, wherein said switching power supply is composed of said power factor correction circuit and a converter, said circuit comprising: a rectifier; a first winding connected with said rectifier; two switch devices connected in series and having a first common node and a first non-grounded end; two diodes respectively connected in anti-parallel with said two switch devices; a second winding connected with said first common node; two capacitors connected in series and having a second common node and a second non-grounded end, wherein said second common node is connected to said first common node through said second winding and said second non-grounded end is connected with said first non-grounded end directly; and a third capacitor connected with said first non-grounded and said second non-grounded end.
 2. The power factor correction circuit of claim 1, wherein said circuit further comprises a control circuit to control the switching state of said two switch devices.
 3. The power factor correction circuit of claim 2, wherein said control circuit is the control circuit of said converter.
 4. The power factor correction circuit of claim 1, wherein said two, switch devices are the switch devices of said converter.
 5. The power factor correction circuit of claim 1, wherein the first winding and the second winding both are primary windings of a transformer of said converter.
 6. The power factor correction circuit of claim 1, wherein said two diodes conduct in a same direction.
 7. The power factor correction circuit of claim 1, wherein said converter is a half-bridge converter.
 8. The power factor correction circuit of claim 1, wherein said first winding is replaced by an independent inductor.
 9. The power factor correction circuit of claim 1, wherein said switch device is a transistor.
 10. A power factor correction circuit for improving a power factor of a switching power supply, wherein said switching power supply is composed of said power factor correction circuit and a converter, said circuit comprising: a rectifier; a first winding connected with said rectifier; two switch devices connected in series and having a common node and a non-grounded end; two diodes respectively connected in anti-parallel with said two switch devices, a second winding connected with said common node; a first capacitor connected to said common node through said second winding; and a second capacitor connected with said non-grounded end.
 11. The power factor correction circuit of claim 10, wherein said circuit further comprises a control circuit to control the switching state of said two switch devices.
 12. The power factor correction circuit of claim 11, wherein said control circuit is the control circuit of said converter.
 13. The power factor correction circuit of claim 10, wherein said two switch devices are switch devices of said converter.
 14. The power factor correction circuit of claim 10, wherein the first winding and the second winding both are primary windings of a transformer of said converter.
 15. The power factor correction circuit of claim 10, wherein said two diodes conduct in a same direction.
 16. The power factor correction circuit of claim 10, wherein said converter is a half-bridge converter.
 17. The power factor correction circuit of claim 10, wherein said first winding is replaced by an independent inductor.
 18. The power factor correction circuit of claim 10, wherein said switch device is a transistor.
 19. A power factor correction circuit for improving a power factor of a switching power supply, wherein said switching power supply is composed of said power factor correction circuit and a converter, said circuit comprising: a rectifier; a first winding connected with said rectifier; first and second switch devices connected in series and having a first common node and a first non-grounded end; first and second diodes respectively connected in anti-parallel with said first and second switch devices; a second winding connected with said first common node; third and fourth switch devices connected in series and having a second common node and a second non-grounded end, wherein said second common node is connected to said first common node through said second winding and said second non-grounded end is connected with said first non-grounded end directly; third and fourth diodes respectively connected in anti-parallel with said third and fourth switch devices; and a first capacitor connected with said first non-grounded end and said second non-grounded end.
 20. The power factor correction circuit of claim 19, wherein said circuit further comprises a control circuit to control the switching state of said first, second, third and fourth switch devices.
 21. The power factor correction circuit of claim 20, wherein said control circuit is a control circuit of said converter.
 22. The power factor correction circuit of claim 19, wherein the first winding and the second winding both are primary windings of a transformer of said converter.
 23. The power factor correction circuit of claim 19, wherein said first and second diodes conduct in a same direction.
 24. The power factor correction circuit of claim 19, wherein said third and fourth diodes conduct in a same direction.
 25. The power factor correction circuit of claim 19, wherein said converter is a full-bridge converter.
 26. The power factor correction circuit of claim 19, wherein said first winding is replaced by an independent inductor. 